Signal acquisition and distance variation measurement system for laser ranging interferometers

ABSTRACT

A signal acquisition and distance variation measurement system for laser ranging interferometers comprising a signal acquisition unit. This unit comprises a frequency detection subsystem to detect the frequency of a received laser beam measurement signal which is buried in noise, wherein the subsystem comprises an acquisition subsystem with a wavelet packet decomposition unit and a phase/frequency detector PLL; a main PLL unit being coupled to the frequency detection subsystem for receiving the detected frequency of the measurement signal for phase estimation of the measurement signal; and a phasemeter band detection subsystem for detecting whether the measurement signal frequency is higher or lower than a reference signal frequency of a reference laser beam by operating a known change to the frequency of the reference signal and measuring the consequent change in the frequency of an interference signal for having the phase of reference signal locked to the measurement signal phase.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims the benefit of the European patent applicationNo. 14003842.3 filed on Nov. 14, 2014, the entire disclosures of whichare incorporated herein by way of reference.

BACKGROUND OF THE INVENTION

The invention relates to a signal acquisition and distance variationmeasurement system for laser ranging interferometers, in particular foruse in a space application, having a signal acquisition unit.

In the past, spaceborne laser terminals have been employed only forcommunication purposes. However, investigations have been made to uselaser instruments as radio instruments due to their reliability. As aresult, future planned missions such as eLISA, NGO and, next generationgeodesy missions are going to implement laser technology in order tomeasure distance variations between satellites of a satelliteconstellation as the interference of a reference beam with a measurementbeam allows performing range measurements with a precision equal to afraction of the laser wavelength. The range measurement relies on atacking phasemeter and on the accuracy with which it is able to evaluatethe phase of the interference signal.

By means of a satellite constellation, gravitational waves and planetgravimetry can be studied, as their orbital trajectories are influencedby these physical phenomena. A technique used to evaluate trajectoryanomalies introduced by gravimetry or by gravitational waves ismeasuring the phase difference of an electromagnetic signal between atransmitting-receiving-satellite. This technique has been employed bythe Gravity Recovery And Climate Experiment (GRACE) which addresses itsmeasurements through a microwave link between a trailing and a followingsatellite. It is known that measurement accuracy can be highly improvedusing a laser link rather than a microwave link.

The accuracy of the laser range measurement relies on a phasemeter,mainly on a PLL (Phase-Locked-Loop), which measures the phase differencebetween a measurement beam and a reference beam. The main architectureof phasemeters is based on so-called mixing phase detector PLLs as theyexploit the most accurate phase measurement when compared to other PLLconcepts. The drawback of these PLLs is the small pull-in-ranges whichconstrains the maximum frequency offset allowable between the inputsignal and the VCO (Voltage Controlled Oscillator).

Before establishing a measurement link, the satellite constellation hasto undergo an initial signal acquisition phase. Independently from thecomplexity of this mandatory process, which can vary according to themission and/or instrument layout, the initial frequency offset betweenthe input signal and the VCO can reach up to 200 MHz or more. Thephasemeter requires a build-in frequency estimation stage algorithm,respectively, in order to properly steer the VCO frequency of the mainPLL towards the incoming beatnote.

From prior art, the importance of the main PLL in addressing the rangemeasurement accuracy is known. In this sense, efforts have been made indeveloping phasemeter prototypes and signal processing algorithms inorder to fulfill a specific mission requirement.

A more broadband view on signal processing for the initial signalacquisition may be made by using FFT (Fast Fourier Transformation) orDFFT (Digital Fast Fourier Transformation) based algorithms. Thesealgorithms, working in a frequency domain, require, in addition to themain PLL board, processing units where FFT algorithms are implemented.These are used to initialize the main PLL and lock the incomingmeasurement signal (Laser Light Signal). FFT based algorithms can alsobe directly implemented in FPGAs (Field Programmable Gate Array).Nevertheless, the amount of FPGA logic elements and memory required bythese algorithms in order to detect with sufficient accuracy the correctfrequency of the received signal in a reasonable amount of time makes italmost impossible to implement a full phasemeter in just a single FPGA.

The inference of the phasemeter detection band is implicitly designatedvia the mandatory signal acquisition phase. Until now, no dedicatedautonomous process for the inference of the phasemeter detection band isknown.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a distance variationmeasurement system for laser ranging interferometers which is improvedover the prior art solutions.

According to the invention, a signal acquisition and distance variationmeasurement system for laser ranging interferometers, in particular foruse in the space application, having a signal acquisition unit isprovided. The signal acquisition unit comprises a frequency detectionsubsystem for detecting the frequency of a measurement signal of areceived laser beam which is buried in noise wherein the frequencydetection subsystem comprises an acquisition subsystem with a waveletpacket decomposition unit and a phase/frequency detector PLL; a main PLLunit being coupled to the frequency detection subsystem for receivingthe detecting frequency of the measurement signal for phase estimationof the measurement signal; a phasemeter band detection subsystem fordetecting whether the frequency of the measurement signal is higher orlower than the frequency of the reference signal of the reference laserbeam by operating a known change to the frequency of the referencesignal and measuring the consequent change in the frequency of aninterference signal for having the phase of the reference signal lockedto the phase of the measurement signal.

The distance variation measurement system according to the inventioncovers the gap of having a discrete time signal processing systemcapable of properly tuning the main PLL VCO frequency and lock theincoming light signal without the use of FFT based algorithms. Thesignal processing system operates in the (discrete) time domain and canbe directly coded with a main PLL in FPGA boards as the general usage ofthe FPGA logic elements and memory is significantly reduced whencompared to FFT based algorithms designed for the same purpose.

The phasemeter band detection subsystem allows a loosening of theconstraint on how to perform the initial signal acquisition phase.

According to a preferred embodiment the acquisition subsystem maycomprise a signal power estimation unit.

According to a further preferred embodiment the acquisition subsystemmay comprise an upsampling unit.

According to a further preferred embodiment, the upsampling unit and allunits after the upsampling unit are not triggered by the wavelet packetdecomposition unit and the power estimation unit in case no measurementsignal is received. This results in a lower probability of faultsdetection.

According to a further preferred embodiment the phasemeter banddetection subsystem may comprise a signal storing unit, a frequencytuning unit and a band detection unit.

According to a further preferred embodiment the main PLL unit maycomprise a mixing-phase detector PLL.

According to a further preferred embodiment the SNR may reach up to −10dB in a frequency band of the measurement signal which spans from 2 MHzto 20 MHz. However, this is not a limiting case. After a proper tuningof the system's parameters and by adding (or removing) stages to thewavelet packet decomposition unit the frequency range of the measurementsignal and SNR can differ from the one reported beforehand.

The distance variation measurement system for laser ranginginterferometers comprises a digital and automated signal acquisitionunit that can operate with low SNRs (Signal-to-Noise-Ratio) of the inputsignal, in particular in a frequency band that spans from 2 MHz to 20MHz where the useful signal reaches up to −10 dB. The signal acquisitionunit is made up of three major subsystems. The frequency detectionsubsystem, the main PLL unit and the phasemeter band detectionsubsystem.

The frequency detection subsystem is used to detect the frequency of themeasurement signal and feed this information to the main PLL subsystem.The frequency detection subsystem combines a wavelet packetdecomposition with a phase/frequency detector PLL.

The main PLL is used to exploit the measurement, i.e., the phaseestimation of the measurement signal, and is implemented usingmixing-phase detector PLL architecture.

The phasemeter band detection subsystem is used to detect whether thefrequency of the received laser beam is higher or lower than thefrequency of the reference laser beam. This information is inferred byoperating a known change to the frequency of the reference beam andmeasuring the consequent change in the frequency of the interferencesignal. This operation is necessary to phase-lock the reference laser tothe received light signal.

In an alternative, the phasemeter band detection subsystem may beomitted if the laser used to generate the reference beam is locked to areference cavity and not phase-locked to the received beam.

The distance variation measurement system according to the invention isconceived for space applications with lasers. However, the distancevariation measurement system described in this application may be usedfor other applications that differ from laser interferometry (ground orspace based) as well. The distance variation measurement system may beused for acquiring and locking a more general electromagnetic signalwhich is buried in noise.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described more detailed by reference to theaccompanying figures. In these figures like reference numerals denotelike elements.

FIG. 1 is a block diagram showing the operating principle of a digitalsignal acquisition unit of a distance variation measurement systemaccording to the invention.

FIG. 2 is a more detailed block diagram of the frequency detectionsubsystem.

FIG. 3 is a more detailed block diagram of a phase/frequency detectorPLL subsystem architecture.

FIG. 4 is a more detailed block diagram of a main PLL unit.

FIG. 5 is a more detailed block diagram of a band detection subsystem.

FIG. 6 is a structure of the second order allpass filter used in the nstage of a wavelet bracket decomposition unit of the frequency detectionsubsystem of FIG. 2.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The phasemeter band detection subsystem allows a loosening of theconstraint on how to perform the initial signal acquisition phase.

FIG. 1 shows a block diagram illustrating the three major subsystems ofthe automated signal acquisition unit of a distance variationmeasurement system according to the invention. The three majorsubsystems are the frequency detection subsystem 100, a main PLL unit200 and a phasemeter band detection subsystem 300.

The frequency detection subsystem 300 is divided into two divisionalsubsystems: a wavelet subsystem 110 and a phase/frequency detector 120.The wavelet subsystem 110 is made up of a wavelet packet decompositionunit 1, a signal power estimation unit 2 and an upsampling unit 3. Amore detailed block diagram of the wavelet subsystem 110 is shown inFIG. 2. A digitalized measurement signal MS is fed into the waveletpacket decomposition unit 1 of the wavelet subsystem 110. The waveletpacket decomposition unit 1 cuts the electronics' frequency range intoseveral sub-bands. The signal in a sub-band has a higher SNR(Signal-to-Noise-Ratio) with respect to the SNR of the signal in theentire frequency range. Each stage 42-1, . . . , 42-n of the waveletpacket decomposition unit 1 doubles the amount of sub-bands and thusincreases the SNR by 3 dB. By means of the wavelet packet decompositionunit 1, the system can operate with a low SNR of the input signal (in afrequency band that spans from 2 MHz to 20 MHz it can reach up to −10dB). The output signals of the wavelet packet decomposition unit 1 arediscrete time signals with a lower sample rate with respect to the inputsignal.

A possible implementation of the wavelet packet decomposition unit 1requires separate filters in each path of its tree structure. Availingof the downsampling, the filters HPF (Highpass Filter) and LPF (LowpassFilter) starting from the second stage 42-2 run with a lower clockfrequency. Taking for simplicity the second stage 42-2 as example,instead of outputting a sample every two, a filter HPF, LPF in thisstage can alternate between the output signals after first stagefilters. The filter implementation of the wavelet packet decompositionunit 1 is better depicted in FIG. 2. Using this expedient for all thestages 42-1, . . . , 42-n of the wavelet packet decomposition unit 1 anarbitrary number n of stages is implemented with only two filters HPF,LPF per stage. The signals between two filter stages are properly routedthrough a multiplexer section signal Sel1, . . . , Seln which isgenerated using the bits of a binary counter 17. Multiplexers of thestages 42-1, . . . , 42-n are denoted with 43. The implementation of thewavelet packet decomposition unit 1 also differs from its classicalimplementation as it omits a last downsampling process. Second orderallpass filters in a minimum multiplierfilter structure are used toimplement the wavelet packet decomposition unit 1. This is depicted inFIG. 6. In the first stage the filter is a common allpass filter andconsists of adders 44, multipliers 46 and registers 47 (z−1). In thesecond stage 42-2, the register outputs of each filter have to match thealternating input signal. Therefore, they need to be doubled together toa shift register. The shift register of the nth stage filter is alwaystwice the size of the shift register of the nth-1 stage filter.

The output values of the wavelet packet decomposition unit 1 are notordered from the lowest to the highest frequency sub-band due to thesignal multiplexing and the implementation of pipelined filters.Pipelining is necessary to allow higher clock frequencies without a hugeconsumption of logic elements. After a first stage 42-1 of the waveletpacket decomposition unit 1 there is always a delay between the inputsignal and the corresponding filtered signal. The task of matching thefilter signal output at every clock cycle with its correspondingsub-band index is addressed by a LUT (Look Up Table) 19 in the signalpower estimation unit 2.

After the wavelet packet decomposition unit 1, the presence of ameasurement signal MS in any of the frequency sub-band is evaluated withthe signal power estimation unit 2. The presence of a measurement signalis detected comparing the mean power values of all sub-bands. The signalpower estimation unit 2 comprises two power estimation components 18.The power estimation components 18 perform power estimation on theoutputs of the wavelet packet decomposition unit 1 in parallel.Initially, the output values of the wavelet packet decomposition unit 1are squared and processed with a first order lowpass filter circuit. Thetotal amount of signal processed by each output of the wavelet packetdecomposition unit 1 equals half the total amount of sub-bands. Ashift-register 20 with the same depth as the signals processed by asignal output of the wavelet packet decomposition unit 1 is used tostore the estimated signal power of each sub-band. Each clock cycle anew output value from the wavelet packet decomposition unit 1 arrives tothe signal power estimation unit 2 and passes through the lowpass filtercircuit. The mean signal power of this sub-band is afterwards comparedwith the value stored in memory 21 which represents the highest meansignal value of all the sub-bands processed in the path. The register 20does not store the highest calculated power value of a sub-band but isalways updated with the most recent value. In this way, the comparisonbetween the latest signal power values of the sub-bands prevents thesystem from not detecting a change in frequency and sub-band of themeasurement signal MS. The time stamp of the signal arrival at powerestimation components 18, which univocally identifies the sub-band beingprocessed by the wavelet packet decomposition unit 1 is evaluated usingthe binary counter 17. The time stamp of the sub-band with the highestsignal power is stored in memory 22 if the sub-band is the result oflowpass filtering process in the wavelet packet decomposition unit 1 orin a memory 23 if the sub-band is the result of highpass filteringprocess in the wavelet packet decomposition unit 1.

For some applications it may be necessary to avoid the output of thelowest frequency sub-bands as small offsets in the ADC(Analog-Digital-Converter) samples increase the power of the noise floorof these sub-bands hiding the measurement signal. For this reason theLUT 19 is implemented in order to exclude these sub-bands. The number oflow frequency sub-bands avoided is related to the specific application.

The last segment of the signal power estimation unit 2 is used tocompare the power values which are stored in memory or register of thetwo sub-bands whose index is stored in the memories 22, 23 respectively.This comparison is performed in comparator 24 and it is used to evaluatein which of the two outputs of the wavelet packet decomposition unit 1the measurement signal MS is located. The outcome of this comparison isused to tune the multiplexing block 26 in the upsampling unit 3, whereonly one of the two parallel outputs of the wavelet packet decompositionunit 1 is processed. In addition, the corresponding register 22 or 23returns the time stamp in which the sub-band with the highest power isprocessed in the wavelet packet decomposition unit 1 and feeds thisinformation to the signal router 27. Through the signal router 27 onlythe sub-band with the highest output power is processed in theupsampling unit 3. The time stamp is also used to recover the propersub-index by means of a look-up table 25.

The output signals of the wavelet packet decomposition unit 1 havefrequency components that span from zero to the new Nyquist frequencywhich is lowered during downsampling. Due to the downsampling andfiltering cascade frequencies that are located in any of the originalsignal spectrum can end up being close to the 0 frequency or to the newNyquist frequency. Generally, phase/frequency detector PLLs havedifficulties in locking a signal if its frequency is too close to the 0frequency or to the Nyquist frequency. Moreover, the reduced clockfrequency of the phase/frequency detector PLL would increase the timerequired to lock the signal.

These system limitations are overcome by the upsampling unit 3. Theupsampling unit 3 does not reconstruct the original signal spectrum butthe frequency components are mapped into a specific frequency rangeregardless of the sub-band where the measurement signal MS is. Thisallows a unique tuning of parameters of the phase detector 4 and theloop filter 5 regardless of the initial frequency of the measurementsignal MS. With a free staged upsampling sequence the upsampling unit 3converts the output of the wavelet packet decomposition unit 1 into asignal whose sampling frequency matches the sampling frequency of themeasurement signal MS before the wavelet packet decomposition unit 1.

As the last downsampling stage is omitted in the implementation of thewavelet packet decomposition unit 1, the output signal of a lowpassfilter in the last stage of the wavelet packet decomposition unit 1 onlycontains frequency components below half of its Nyquist frequency whilethe output signal of the highpass filter contains frequency componentsaround the Nyquist frequency. In the upsampling unit 3 after the element27, the signal is then upsampled by a factor of two. The upsampling isperformed inserting additional zeros (so-called zero padding technique)between each sampled value. This technique introduces periodicrepetitions of the signal frequency components in the frequencyspectrum. After this first upsampling process, frequency componentsclose to the zero a mirrored close to the new Nyquist frequency (and allits multiples) while any high frequency component remains close to theold Nyquist frequency (and all its multiples). For such a reason,instead of implementing an anti-imaging lowpass filter in the order tocut a higher frequency component a highpass filter 28 having the samestructure as a high pass filter of the wavelet packet decomposition unit1 is used. The filtering removes any frequency component close to zerowhile any higher frequency component or mirrored frequencies are notfiltered. Changes in the amplitude of the signal are avoided using again amplification of two.

In the second stage of the upsampling unit 3 the signal is once moreupsampled and then filtered using a lowpass filter 29. This time, thefiltering is performed using a lowpass filter 29 in order to remove allfrequency components that are close or higher than the Nyquist frequencyafter this new upsampling. The frequency response of the filter has tobe very sharp in order to avoid signal attenuation in the frequencyrange of interest. For this filtering process, a lowpass filterstructure as the one implemented in the wavelet packet decompositionunit 1 is used.

The original sampling frequency is achieved by means of a lastupsampling stage 30. Since the repetitions in the frequency spectrum arewidely separated, a single lowpass filter 31 is used to perform theanti-imaging. In this case a fifth order elliptic lowpass filter in theminimum multiplier allpass structure is used. Regardless of the totalamount of stages in the wavelet packet decomposition unit 1 after theupsampling unit 3 all frequency components are mapped into the third orfourth sub-band in which the wavelet packet decomposition unit 1 hassubdivided the frequency spectrum (i.e., the sub-band location of themeasurement signal is related to the last filtering process of 1 thatlocates the frequency of the measurement signal to its sub-band).

Both, the wavelet packet decomposition unit 1 and the upsampling unit 3,need sharp halfband filters. For a lowpass filter, the attenuation ofthe higher frequencies has to minimize the impact on the remainingsignal through aliasing. The attenuation in the passband must be smallenough so that signals in the whole remaining frequency range can bedetected. To test the system, the filters implemented in the waveletpacket decomposition unit 1 and the upsampling unit 3 are 13th orderelliptic filters with a maximum passband ripple of 0.5 dB. Even thoughthis attenuation causes a signal power loss of almost 11% it does notcompromise the signal detection. The passband edge is chosen to be atnormalized frequency 0.25 and the stopband starts at 0.25005. In thestopband the attenuation is higher than 34 dB. Thus, the noise power ofthe measurement signal is reduced of 3 dB. The values used in theprevious description are just an example of a possible filterimplementation. A different implementation would change the systemperformance but not the operating principle.

The outcome of the upsampling unit 3 is fed into a phase detector unit 4of the phase/frequency detector PLL 120. A more detailed block diagramof the phase/frequency detector PLL 120 and the phase detector unit 4 isdepicted in FIG. 3. The phase detector unit 4 which is used to determinedifferences in phase and frequency by comparing the rising edges of theoscillator 6 and the input from the upsampling unit 3 using twoD-flipflops 32. For simplicity, it is possible to represent both theedge detection signals each with a single bit. A rising edge is detectedstoring the previous sample value and comparing it with the actualvalue. If a rising edge is detected from the upsampling unit 3, theauxiliary signal vu becomes one. If a rising edge occurs in theoscillator 6, a signal vd is set to one. In case both vu and vd areequal to one, both D-flipflops 32 are reset. In the case only one of thetwo signals is unitary, the charge pump's (33) output value is increasedif vu is unitary or decreased if vd is unitary. The digitalimplementation of the charge pump 33 is a limited integrator whoseabsolute value is decreased at each clock cycle by the factor Ka. Forlock detection, the absolute value of the phase detector unit 4 is fedinto a lowpass filter 35. The parameters of a loop filter 5 areoptimized in order to reduce the lock time of this PLL. The frequencyvalue fed to the oscillator 6 is directly the output value of the loopfilter 5. In the locked state the output value of the loop filter isclose to zero. Therefore, the loop filter 5 has to include anintegrator. The loop filter 5 is implemented using a PI controller.

Since the two input signals of the phase detector unit 4 are singlebits, the oscillator 6 consists of a phase register that accumulates afrequency value at its input. The output signal of the oscillator 6 isthe MSB (Most Significant Bit) of the phase register. Since theoscillator 6 is implemented in digital hardware the range of 0 to 2π ismapped to a power of two such that overflows take care of the modulooperation. The maximum phase step in one clock cycle is π and equalshalf of the clock frequency. Larger values would lead to a lower outputfrequency and so the frequency word can be one bit smaller than thephase accumulator. The register length of the oscillator 6 can beparameterized in order to make an adequate frequency guess. With a clockfrequency of 100 MHz, a phase accumulator with a register length of 16bits returns in a minimum frequency step of 1526 Hz. This value issufficiently small and allows the main PLL to 7 (cf. FIG. 1) to pull-inand lock the input signal. The register size is incremented in presenceof high SNR in order to avoid the loss of the LSBs (Least SignificantBits).

In order to detect whether the lock frequency of the phase/frequencydetector PLL 120 is reliable, the output of the register 34 is processedthrough the lowpass filter 35. The absolute value of the register 34 isfed into the lowpass filter 35 which returns the first four MSBs of itsfiltered signal. The lowpass filter 35 is used as a performanceindicator that controls the re-initialization of the main PLL frequencythrough the main PLL 7. As the signal processed in the lowpass filter 35is already smoothed by the charge pump 33, the filter implemented is afirst order Butterworth filter.

The oscillator frequency of the phase/frequency detector PLL 120 whichis used for the frequency estimation in the frequency recovery unit 7 isnot derived directly from the oscillator 6 but as outcome of the loopfilter 5. The frequency is estimated with loop filter 5 only using theintegral part 36 of the PI controller due to the behavior of thephase/frequency detector PLL 120 in a noisy environment. The integralpart 36 tunes the oscillator in order to match the frequency of theinput signal while the proportional part of the loop filter compensatesthe residual phase difference. Since this PLL structure is sensitive tocycle slips in presence of noise, the input of the oscillator 6 is usedto track the phase. Therefore, its mean frequency can differ from thefrequency of the PLL's input signal. The integral part 36 is instead amore accurate estimate of the frequency of the PLL's input signal.

The frequency of the measurement signal is estimated in the main PLL 7based on the outcome of look-up-table (LUT) 25 and the loop filter 5.Since the sub-band of the wavelet packet decomposition unit 1 with thehighest signal power is not reconstructed but upsampled, the frequencyvalue of loop filter 5 cannot be fed directly into the loop filter 10which is part of the main PLL unit 200. The frequencies in odd numberedsub-bands of the wavelet packet decomposition unit 1 (starting to countfrom low frequencies to high frequencies and labelling as zero the firstsub-band) are mirrored when using the classical implementation of thewavelet packet decomposition unit 1 which includes a final signaldownsampling. Being this not the case of the chosen implementationarchitecture, the output signals of the highpass filter remain above theNyquist frequency and are not mirrored. In terms of sub-bands thisresults in the two lowest sub-bands not being mirrored, the followingtwo being mirrored and so forth. During the upsampling process in theupsampling unit 3, all of the frequencies are mirrored regardless on thesub-band. As mentioned beforehand, the upsampling process is designedsuch that the outcome of a highpass filter is mirrored in the thirdsub-band in which the wavelet packet decomposition unit 1 has dividedthe electronics frequency range while the outcome of the lowpass filteris always mirrored in the fourth sub-band of this frequency range. Thisexpedient allows optimizing the performance of the phase/frequencydetector PLL 120 around a well-defined frequency interval.

In order to recover the original frequency value of the input signal,the two LSBs of the sub-band index (also expressed in bits) derived inthe signal power estimation unit 2 from the look-up-table 25 are used.This leads to four different cases:

LSBs=00: The frequencies are mirrored after the upsampling unit 3. Thefrequency sub-band containing the input signal is obtained after alowpass filtering process. Due to the upsampling unit 3, all frequencycomponents can be found in the fourth sub-band of the electronicsfrequency range. The input frequency of the loop filter 10 is(isb+4)·Δfsb−fosc being isb is the sub-band index expressed in decimalunits derived from the look-up-table 25, Δfsb is the frequency width ofthe sub-bands and fosc is the frequency value derived from the output ofthe integral part 36 in the loop filter 5.

LSBs=01: The frequencies are mirrored after the upsampling unit 3. Thefrequency sub-band containing the input signal is obtained after ahighpass filtering process. Due to the upsampling unit 3, all frequencycomponents can be found in the third sub-band of the frequency spectrum.The input frequency of the loop filter 10 is (isb+3)·Δfsb−fosc where theequation terms have been defined beforehand.

LSBs=10: The frequencies are not mirrored after the upsampling unit 3.The frequency sub-band containing the input signal is obtained after ahighpass filtering process. Due to the upsampling unit 3, all frequencycomponents can be found in the third sub-band of the frequency spectrum.The input frequency of the loop filter is (isb−2)·Δfsb+fosc where theequation terms have been defined beforehand.

LSBs=11: The frequencies are not mirrored after the upsampling unit 3.The frequency sub-band containing the input signal is obtained after alowpass filtering process. Due to the upsampling unit 3, all frequencycomponents can be found in the fourth sub-band of the frequencyspectrum. The input frequency of the loop filter 10 is (isb−3)·Δfsb+foscwhere the equation terms have been defined beforehand.

The output of lowpass filter 35 is used to decide whether the frequencyof the main PLL unit 200 has to be reinitialized. The estimatedfrequency is compared to the frequency value of the main PLL 200measured in the register 37 of loop filter 10 (see FIG. 4). If theirdifference exceeds a prefixed value (defined in a LUT also implementedin the frequency recovery unit 7) and the main PLL is not alreadylocked, its frequency is reinitialized. Since the lowpass filter 35 isan index of the reliability of the frequency estimation (lower valuesare better) the first bit of the lowpass filter 35 has to be zero. There-initialization occurs by setting the frequency value in the register37 to the one determined with the criteria described beforehand. Inorder to feed the frequency estimated in the frequency recovery unit 7in the register 37 of the loop filter 10 of the main PLL unit 200,additional zeros have to be added as LSBs as the register size of theregister 37 is longer than the frequency bit word calculated in thefrequency recovery unit 7. The main PLL frequency re-initialization isoperated in the register 37 so that the oscillator 11 of main PLL unit200 can directly process the output signal of the loop filter 10. Oncethe main PLL unit 200 reaches a locked state, a new initialization ofthe register 37 in the loop filter 10 occurs only if the main PLL unit200 loses lock.

A detailed block diagram of the main PLL unit 200 is depicted in FIG. 4.Its implementation is based on its classical architecture and thereforecomprises a multiplier 8, a lowpass filter 9, the loop filter 10 and anoscillator 11. The phase difference calculation between the measurementsignal MS and the output signal of 11 is performed by multiplying, foreach sample, the measurement signal MS with the cosine of the signalgenerated in 11. The high frequency component that derives from thisproduct has to be attenuated with influencing the loop performance. Forthis purpose an IIR Butterworth lowpass filter 9 with a very lownormalized cut of frequency is implemented. The architecture of thelowpass filter 9 consists in first order filters working in cascade anduses powers of two as filter coefficients in order to simplify theoverall filter structure. The cut-off frequency can be changed by eithertuning the coefficients or varying the filter order. It is important notto truncate intermediate results as well as the outputs of a filteringprocess but to round them. Otherwise the resulting additive errorprocessed in the loop filter would have a negative mean value. Thiswould worsen the PLL performance in presence of a large dynamic range ofthe measurement signal as the constant magnitude of the error signalcauses variable phase offsets that depends on the amplitude of themeasurement signal.

The error signal used to operate the loop filter (which is derived afterthe lowpass filter 9) cannot be used as a lock detection flag. If thePLL is locked the error signal is small and its dynamics is driven bythe noise of the measurement signal MS. If the two frequencies have anoffset in the range of the loop bandwidth, the error signal amplitudeincreases.

Nevertheless, a small amplitude of the error signal is also measuredwhen the frequency of the oscillator 11 and the frequency of themeasurement signal MS have a big offset. This artefact is caused by thelowpass filter 9 as it attenuates the higher frequency components. Inorder to evaluate if the PLL is locked or not, the measurement signal isdirectly mixed with the second output of the oscillator 11 as well. Ifthe PLL is close to locking the measurement signal MS, the mean value ofthis product equals approximately half of the amplitude of themeasurement signal otherwise it is a duplicate of the error signal. Alock is achieved when the ratio between these two signals exceeds afixed threshold value (the magnitude of this threshold value is relatedto the noise floor and the maximum amplitude of the measurement signal).As a consequence of the lowpass filtering process in this calculation,the lock signal is delayed of approximately 50 μs in time.

The implementation of the loop filter 10 is the same as in the loopfilter 5 with a difference in the controller parameters and registerssize (larger registers are used for the loop filter 10 in order toperform more accurate measurements). The parameters of the oscillator 11are chosen in order to guarantee optimal performance in terms of phasemeasurements and locking time. They can be evaluated using the classicaltheory of PI controllers.

A loop bandwidth of 10 kHz increases the phase measurement accuracywhile a loop bandwidth of 100 kHz reduces the time required from themain PLL to lock the measurement signal. For such a reason, the main PLLis designed to operate in two different modes: acquisition mode with 100kHz loop bandwidth and measurement mode with 10 kHz loop bandwidth,respectively. When the main PLL unit 200 is locked to the measurementsignal MS, it is operated in measurement mode otherwise in acquisitionmode. The mode switch is controlled by monitoring the lock signal and isoperated by changing the parameters of the lowpass filter 9 and the loopfilter 10.

The loop filter 10 locks to the measurement signal if the output signalof the oscillator 11 is phase shifted of −90 degrees with respect to themeasurement signal MS. For the lock detection, a sinusoidal output isalso implemented in the oscillator 11. The architecture of theoscillator 11 uses a single phase accumulator 38 and a sine table bymeans of a dual port ROM 39. Due to the symmetrical properties of asinusoidal signal, it is sufficient to store only the values of aquarter of a period plus one.

For the high precision phase measurements it is necessary that the phaseaccumulator has no overflow at 2π. On the other side it is hardlypossible to implement huge look-up-tables in embedded hardware. For thisreason only some of the middle bits are used to address thelook-up-table and the MSBs and LSBs are discarded. The register lengthof the phase accumulator and the frequency word can be independentlyparameterized.

The last stage of the phasemeter is dedicated to the decimation of thedata of interest for consequent signal processing. Since the frequencycomponents in the signal of interest are below one Hz, extensivedownsampling can be performed. In the downsampling process ananti-aliasing lowpass filter is required. A downsampling factor of 222decreases the sampling frequency from 50 MHz to about 12 Hz. To avoidaliasing artefacts in the signal of interest the attenuation around theoutput sampling frequency at 11 Hz to 13 Hz has to be sufficiently high.If aliasing artefacts in the higher frequency range are allowed, anaccumulator can take care of the filtering. The accumulator adds theinput signal values to its register until a rising edge in the sloweroutput clock signal occurs. At that point in time the decimation filterprovides the current accumulator value on its output while theaccumulator itself is reset to zero. Basically this is a FIR filter witha rectangular impulse response. In the frequency domain this correspondsto a sin c function. Even if the attenuation at higher frequencies isvery low and leads to heavy aliasing, this filter has a huge advantage.The zeros of the sin c function are located at every multiple of theoutput sampling frequency. This ensures that the signal of interest isnot affected by aliasing and thus a good reason for using this verysimple filter structure.

Once the main PLL unit 200 is locked and has switched to measurementmode, the lock signal and the lock frequency are sent to the BandDetection Subsystem 300. A detailed block diagram of this subsystem isdepicted in FIG. 5. Here, the lock frequency measured in the loop filter10 is stored in a register 13. The lock signal is also used to trigger aramp signal 40 in a frequency tuning unit 14 which is continuously addedto the lock frequency value stored in the register 13. The ramp signal40 is used to change the laser frequency in a well-defined way. Sincethe main PLL unit 200 in measurement mode is capable of detectingfrequency changes in the KHz region, the ramp is used to change thefrequency of the laser of a few hundred of KHz. The exact value isrelated to the signal noise floor and the sensitivity of the main PLLunit 200. A loop filter 16 is used to operate a PZT of the lasercrystal. Therefore, the ramp signal 40 requires a slope compatible witha maximum frequency change rate achievable with the PZT. The updatedfrequency value is compared with the frequency lock value stored in theregister 13 and determines the detection band using a look-up-table 41in the band detection unit 15. The two possible outcomes of thecomparison determine the detection band of the phasemeter. Inparticular, if the frequency during the ramp tuning increases withrespect to the frequency stored in the register 13, the frequency of thereference laser beam is higher than the frequency of the received beam.In the opposite case, the frequency decreases during the ramp tuning.This information is used to initialize properly the parameters of thelaser control unit and lock the laser to the incoming light signal. Awrong assumption on the phasemeter detection band would lead to anerroneous tuning of the laser frequency while trying to lock it to theincoming light signal. This would rapidly bring the whole phasemeter outof lock. Once the phasemeter band has been detected, the Band Detectionsubsystem 300 is bypassed and the laser frequency is controlled by itscontrol unit.

While at least one exemplary embodiment of the present invention(s) isdisclosed herein, it should be understood that modifications,substitutions and alternatives may be apparent to one of ordinary skillin the art and can be made without departing from the scope of thisdisclosure. This disclosure is intended to cover any adaptations orvariations of the exemplary embodiment(s). In addition, in thisdisclosure, the terms “comprise” or “comprising” do not exclude otherelements or steps, the terms “a” or “one” do not exclude a pluralnumber, and the term “or” means either or both. Furthermore,characteristics or steps which have been described may also be used incombination with other characteristics or steps and in any order unlessthe disclosure or context suggests otherwise. This disclosure herebyincorporates by reference the complete disclosure of any patent orapplication from which it claims benefit or priority.

REFERENCE LIST

-   1 wavelet packet decomposition unit-   2 signal power estimation unit-   3 upsampling unit-   4 phase detector unit-   5 loop filter-   6 oscillator-   7 frequency recovery unit-   8 multiplier-   9 lowpass filter (Butterworth filter)-   10 loop filter-   11 oscillator-   12 decimation-   13 register-   14 frequency tuning unit-   15 band detection unit-   16 loop filter-   17 binary counter-   18 power estimation components-   19 look-up table (LTU)-   20 shift register-   21 memory-   22 memory-   23 memory-   24 comparator-   25 look-up table (LUT)-   26 multiplexing block-   27 signal router-   28 highpass filter-   29 lowpass filter-   30 upsampling stage-   31 lowpass filter-   32 D-flipflop-   33 charge pump-   34 register-   35 lowpass filter-   36 integral part-   37 register-   38 single phase accumulator-   39 ROM-   40 ramp signal-   41 look-up table (LUT)-   42 stage filter-   43 multiplexer-   100 frequency detection subsystem-   110 wavelet subsystem-   120 phase/frequency detector PLL-   200 main PLL unit-   300 phasemeter band detection subsystem-   MS measurement signal-   DF detected frequency-   v_(n) auxiliary signal-   v_(d) signal-   ADC Analog to Digital Converter-   CU Control Unit-   DFFT Digital Fast Fourier Transform-   FFT Fast Fourier Transform-   FPGA Field Programmable Gate Array-   LSB Least Significant Bit-   LUT Look Up Table-   MSB Most Significant Bit-   PI Proportional-Integral-   PLL Phase-Locked Loop-   SNR Signal-to-Noise Ratio-   VCO Voltage Controlled Oscillator-   HPF Highpass Filter-   LPF Lowpass Filter

1. A signal acquisition and distance variation measurement system forlaser ranging interferometers having a signal acquisition unit,comprising: a frequency detection subsystem configured to detect thefrequency of a measurement signal of a received laser beam which isburied in noise, wherein the frequency detection subsystem comprises anacquisition subsystem with a wavelet packet decomposition unit and aphase/frequency detector PLL; a main PLL unit being coupled to thefrequency detection subsystem configured to receive the detectedfrequency of the measurement signal for phase estimation of themeasurement signal; a phasemeter band detection subsystem configured todetect whether the frequency of the measurement signal is higher orlower than the frequency of a reference signal of a reference laser beamby operating a known change to the frequency of the reference signal andmeasuring the consequent change in the frequency of an interferencesignal for having the phase of the reference signal locked to the phaseof the measurement signal.
 2. The signal acquisition and measurementsystem according to claim 1, wherein the acquisition subsystem comprisesa signal power estimation unit.
 3. The signal acquisition andmeasurement system according to claim 1, wherein the acquisitionsubsystem comprises an upsampling unit.
 4. The signal acquisition andmeasurement system according to claim 2, wherein the upsampling unit andall units after the upsampling unit are not triggered by the waveletpacket decomposition unit and the signal power estimation in case nomeasurement signal is received.
 5. The signal acquisition andmeasurement system according to claim 1, wherein the phasemeter banddetection subsystem comprises a signal storing unit, a frequency tuningunit and a band detection unit.
 6. The signal acquisition andmeasurement system according to claim 1, wherein the main PLL unitcomprises a mixing-phase detector PLL.